Noise resistant small signal sensing circuit for a memory device

ABSTRACT

Apparatus and method for data sensing circuitry that uses averaging to sense small differences in signal levels representing data states. The apparatus periodically switches the coupling of input terminals and output terminals of an integrator circuit from a first configuration to a second configuration, where the second configuration changes the polarity of the integrator circuit from the first configuration. The output signals of the integrator circuit are periodically compared, and based on the comparison, output signals having a voltage representative of a value are generated. The values of the output signals are then averaged over time to determine the data state.

TECHNICAL FIELD

The present invention relates generally to integrated circuit memorydevices, and more specifically, to sensing circuitry for sensing smallresistance differences in memory cells, such as in resistive memorycells.

BACKGROUND OF THE INVENTION

Computer systems, video games, electronic appliances, digital cameras,and myriad other electronic devices include memory for storing datarelated to the use and operation of the device. A variety of differentmemory types are utilized in these devices, such as read only memory(ROM), dynamic random access memory (DRAM), static random access memory(SRAM), flash memory (FLASH), and mass storage such as hard disks andCD-ROM or CD-RW drives. Each memory type has characteristics that bettersuit that type to particular applications. For example, DRAM is slowerthan SRAM but is nonetheless utilized as system memory in most computersystems because DRAM is inexpensive and provides high density storage,thus allowing large amounts of data to be stored relatively cheaply. Amemory characteristic that often times determines whether a given typeof memory is suitable for a given application is the volatile nature ofthe storage. Both DRAM and SRAM are volatile forms of data storage,which means the memories require power to retain the stored data. Incontrast, mass storage devices such as hard disks and CD drives arenonvolatile storage devices, meaning the devices retain data even whenpower is removed.

Current mass storage devices are relatively inexpensive and highdensity, providing reliable long term data storage at relatively cheap.Such mass storage devices are, however, physically large and containnumerous moving parts, which reduces the reliability of the devices.Moreover, existing mass storage devices are relatively slow, which slowsthe operation of the computer system or other electronic devicecontaining the mass storage device. As a result, other technologies arebeing developed to provide long term nonvolatile data storage, and,ideally, such technologies would also be fast and cheap enough for usein system memory as well. The use of FLASH, which provides nonvolatilestorage, is increasing in popularity in many electronic devices such asdigital cameras. While FLASH provides nonvolatile storage, FLASH is tooslow for use as system memory and the use of FLASH for mass storage isimpractical, due in part to the duration for which the FLASH canreliably store data as well as limits on the number of times data can bewritten to and read from FLASH.

Due to the nature of existing memory technologies, new technologies arebeing developed to provide high density, high speed, long termnonvolatile data storage. One such technology that offers promise forboth long term mass storage and system memory applications isMagneto-Resistive or Magnetic Random Access Memory (MRAM). FIG. 1 is afunctional diagram showing a portion of a conventional MRAM array 100including a plurality of memory cells 102 arranged in rows and columns.Each memory cell 102 is illustrated functionally as a resistor since thememory cell has either a first or a second resistance depending on amagnetic dipole orientation of the cell, as will be explained in moredetail below. Each memory cell 102 in a respective row is coupled to acorresponding word line WL, and each memory cell in a respective columnis coupled to a corresponding bit line BL. In FIG. 1, the word lines aredesignated WL1-3 and the bit lines designated BL1-4, and may hereafterbe referred to using either these specific designations or generally asword lines WL and bit lines BL. Each of the memory cells 102 storesinformation magnetically in the form of an orientation of a magneticdipole of a material forming the memory cell, with a first orientationof the magnetic dipole corresponding to a logic “1” and a secondorientation of the magnetic dipole corresponding to a logic “0.” Theorientation of the magnetic dipole of each memory cell 102, in turn,determines a resistance of the cell. Accordingly, each memory cell 102has a first resistance when the magnetic dipole has the firstorientation and a second resistance when the magnetic dipole has thesecond orientation. By sensing the resistance of each memory cell 102,the orientation of the magnetic dipole and thereby the logic state ofthe data stored in the memory cell 102 can be determined.

The stored logic state can be detected by measuring the memory cellresistance using Ohm's law. For example, resistance is determined byholding voltage constant across a resistor and measuring, directly orindirectly, the current that flows through the resistor. Note that, forMRAM sensing purposes, the absolute magnitude of resistance need not beknown, the inquiry is whether the resistance is greater or less than avalue that is intermediate to the logic high and logic low states.Sensing the logic state of an MRAM memory element is difficult becausethe technology of the MRAM device imposes multiple constraints. In atypical MRAM device, an element in a high resistance state has aresistance of about 950 kΩ. The differential resistance between a logic“1” and a logic “0” is thus about 50 kΩ, or approximately 5% of scale.

Therefore, there is a need for a sensing circuit for a resistancemeasuring circuit to repeatably and rapidly distinguish resistancevalues for devices having small signal differentials.

SUMMARY OF THE INVENTION

The present invention is directed to an apparatus and method for datasensing that uses averaging to sense small differences in signal levelsrepresenting data states. The apparatus includes an integrator circuithaving a first integrator input electrically coupled to a referencelevel, a second integrator input to which an input is applied, and firstand second integrator outputs at which first and second output signalsare provided, respectively. The integrator circuit further includes aiiamplifier circuit having pairs of differential input and output nodes.The integrator circuit periodically switches the electrical coupling ofeach of the differential input nodes to a respective integrator inputand the electrical coupling of each of the differential output nodes toa respective integrator output. The apparatus further includes acomparator having first and second input nodes electrically coupled to arespective integrator output and further having an output node. Theclocked comparator periodically compares voltage levels of the first andsecond input nodes and generating an output signal having a logic statebased therefrom. A current source having first and second current outputnodes coupled to a respective integrator output is also included in theapparatus. The current source switching the coupling of each currentoutput node to a integrator output based on the logic state of theoutput signal of the comparator.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a functional block diagram showing a portion of a conventionalMRAM array.

FIG. 2 is a functional block diagram of a sensing circuit according toan embodiment of the present invention.

FIG. 3 is a schematic drawing of an integrator stage according to anembodiment of the present invention.

FIG. 4 is a schematic drawing of a switching current source according toan embodiment of the present invention.

FIG. 5 is a schematic drawing of clocked comparator according to anembodiment of the present invention.

FIG. 6 is a functional block diagram of a sensing circuit according toanother embodiment of the present invention.

FIG. 7 is a functional block diagram illustrating an MRAM including asensing circuit according to the present invention.

FIG. 8 is a functional block diagram illustrating a computer systemincluding the MRAM of FIG. 5.

DETAILED DESCRIPTION OF THE INVENTION

Embodiments of the present invention are directed to a noise resistantsensing circuit for data sensing circuitry that uses averaging to sensesmall differences in signal levels representing data states, such as inresistor-based memory circuits. Certain details are set forth below toprovide a sufficient understanding of the invention. However, it will beclear to one skilled in the art that the invention may be practicedwithout these particular details. In other instances, well-knowncircuits, control signals, and timing protocols have not been shown indetail in order to avoid unnecessarily obscuring the invention.

FIG. 2 illustrates an embodiment of a sensing circuit 200 according toan embodiment of the present invention. The sensing circuit 200 includesan integrator stage 210, a switching current source 212, and a clockedcomparator 214. As will be explained in more detail below, an outputsignal UP (or DOWN) of the sensing circuit 200 is averaged over a periodof time to determine the data state stored on a memory cell, such as aresistive memory cell 220. The average value calculated is indicative ofthe data state of the memory cell. In summary, the sensing circuit 200outputs a stream of bits resulting from the cyclical charging anddischarging of capacitors 340, 342. The ratio of logic “1” bits (oralternatively, logic “0” bits) to a total number of bits yields anumerical value that corresponds to an average current through a memorycell, such as resistive memory cell 220, in response to an appliedvoltage. The average current, in turn, is used to determine the logicstate of the data stored by the resistive memory cell 220. Circuitry forperforming the averaging operation of the bit stream provided by thesensing circuit 200 has not been shown or described in great detail inorder to avoid obscuring description of the present invention. A moredetailed explanation of using current averaging for memory cell sensingis provided in the commonly assigned, co-pending U.S. patent applicationSer. No. 09/938,617, filed Aug. 27, 2001, entitled RESISTIVE MEMORYELEMENT SENSING USING AVERAGING, which is incorporated herein byreference.

A potential issue, however, with the circuit described in theaforementioned patent application is related to offset voltages andcurrents inherent with the differential amplifier of the sensingcircuit, as well as 1/ƒ noise. It will be appreciated that these effectscan cause currents in the tens of nano-amperes to be output by thedifferential amplifier. In light of the small voltage margin between twodata states of a memory cell, such as a resistive memory cell, and theresulting magnitude of output current (˜100 nA range) by thedifferential amplifier when reading the memory cell, inherent offsetvoltages and currents, as well as 1/ƒ noise can cause reading errors ifnot compensated. In the sensing circuit described in the aforementionedpatent, offset issues are compensated for by calibrating thedifferential amplifier. However, it is often the case that thecalibration must be adjusted for process variations in fabrication ofthe memory device. Additionally, the process of calibrating thedifferential amplifiers of a memory device is time consuming. As will beexplained in more detail below, embodiments of the present invention,including the arrangement illustrated in FIG. 2, provide offset and 1/ƒnoise compensation without the need for calibration, and allows for theintegration to run indefinitely.

The operation of the sensing circuit 200 will be described generallywith respect to FIG. 2. The resistance Rcell of the resistive memorycell 220 is measured as an input voltage relative to ground. In readinga memory cell, a wordline (WL) 224 corresponding to a row address isactivated and goes HIGH, and bit lines of a memory array are coupled tothe input nodes 226 of the respective sensing circuits 200. All otherwordlines in the memory array are grounded. As illustrated in FIG. 2,the voltage level of the selected WL 224 is dropped over Rcell and a“sneak” resistance 222 that represents the resistance of the otherresistive memory cells of the bit line coupled to the input node 226,but not coupled to the selected WL 224. Note that the ground nodecoupled to the sneak resistance 222 represents the unselected, that is,grounded, wordlines.

For the present example, operation of first and second chopping orswitching circuits 230, 234 will be ignored until later in order tosimplify the explanation of the sensing circuit 200. The voltage appliedto the input node 226 causes a differential amplifier 232 to supplycurrent to either node 236 or 238, and draw current from the other node.As a result, the capacitor coupled to the node to which the differentialamplifier 232 is supplying a current will be charged, increasing thevoltage of the node. Conversely, the capacitor coupled to the node fromwhich the differential amplifier 232 is drawing current will bedischarged, decreasing the voltage of that node. A clocked comparator250 senses the relative voltages of the nodes 236, 238 in response to aclock signal Comp_clk and generates a corresponding output signal UP.The clocked comparator 250 also generates a complementary output signalDOWN. As illustrated in FIG. 2, an inverter 252 is coupled to the outputof the clocked comparator 250 to generate the DOWN signal. However, itwill be appreciated that the clocked comparator 250 is provided by wayof example, and a clocked comparator suitable for use with the presentinvention can be implemented in many different ways other than thatshown in FIG. 2.

The UP and DOWN signals are provided to the switching current source 212having a first current source 260 and a second current source 261. Eachof the current sources 260, 261 switch between being coupled to thenodes 236, 238 based on the state of the UP and DOWN signals. In onestate, the current source 260 is coupled to the node 236, providingcurrent to positively charge the capacitor 236, and the current source261 is coupled to the node 238, providing current to negatively chargethe capacitor 238. In the other state, the current source 260 is coupledto the node 238, providing current to positively charge the capacitor238, and the current source 261 is coupled to the node 236, providingcurrent to negatively charge the capacitor 236. Consequently, where theUP and DOWN signals switch states, the coupling of the current sources260, 261 will switch as well.

For example, as illustrated in FIG. 2, the UP and DOWN signals are LOWand HIGH, respectively, causing the current source 260 to be coupled tothe node 236 and the current source to be coupled to the node 238. Uponthe next rising edge of the Comp_clk signal, the voltages of the nodes236, 238 are sensed by the clocked comparator 250. The voltages at thenodes 236, 238 are represented by signals intout1 p and intout1 m,respectively. Where the coupling of the current sources 260, 261 aresuch that the current provided to the capacitors 240, 242 over theperiod of the Comp_clk signal causes the voltages of the nodes 236, 238to change from the previous rising edge of the Comp_clk signal, theoutput of the clocked comparator 250 changes logic states. This in turncauses the coupling of the current sources 260, 261 to switch nodes aswell. It will be appreciated that the coupling of the current sources260, 261 will continue to switch until the current provided by thedifferential amplifier 232 to either one of the capacitors 240, 242causes the voltage of the respective node 236, 238 to be greater thanthe change in voltage caused by the current source over one period ofthe Comp_clk signal. When this occurs, the logic states of the UP andDOWN signals maintain their present logic states, which causes theaverage of the output signal of the sensing circuit 200 to change.

As previously mentioned, operation of the first and second switchingcircuits 230, 234 has been ignored. Operation of the first and secondswitching circuits 230, 234 will now be discussed. Explained briefly,the first and second switching circuits 230, 234 are used to zero outany inherent offset with the differential amplifier 232 and 1/ƒ noise.As previously discussed, offset voltages and currents, as well as 1/ƒnoise cause reading errors if not compensated. As will be explained inmore detail below, in embodiments of the present invention, offset and1/ƒ noise compensation can be provided without the need for calibration,and additionally, integration can be run indefinitely.

FIG. 3 illustrates an embodiment of an integrator stage 300 that can besubstituted for the integrator stage 210 in FIG. 2. The integrator stage300 includes an input multiplexer 320 coupled to a first switchingcircuit 14 at an input node 322. The multiplexer 320 selects betweencoupling a first digit line signal SA_in_(—)0 and a second digit linesignal SA_in_(—)1 to the input node 322 based on the logic states ofaddress signals B0 and B1. The address signals B0 and B1 areconventional, and provision of these types of signals to the integratorstage 300 are well known in the art. A second input node 324 of thefirst switching circuit 314 is coupled to ground through a transistor328. The gate of the transistor 328 is coupled to a power supply makingthe transistor 328 conductive. Output nodes 332, 334 are coupled tonon-inverting and inverting inputs of the differential amplifier 310,respectively. Non-inverting and inverting outputs of the differentialamplifier 310 are coupled to input nodes 336, 338, respectively, of thesecond switching circuit 318. Output nodes 346, 348 of the secondswitching circuit 318 are coupled to capacitors 340, 342, respectively.As previously discussed, the intout1 p and intout1 m signals provided bythe integrator stage 210 (FIG. 2) to the clocked comparator 214represent the voltages at the nodes 330, 332 as the capacitors 340, 342charge and discharge. The integrator stage 300 further includes acommon-mode feedback circuit 352 coupled to the output nodes 346, 348 ofthe second switching circuit 318 to limit the output current of thedifferential amplifier 310 to a differential current. The voltages,Vbias1, Vbias2, Vbias3, and Vbias4, illustrated in FIG. 3 are biasvoltages that can be generated and provided to the integrator stage 300in any conventional manner. It will be appreciated that selection of thespecific voltage levels can be made by those of ordinary skill in theart based on the description of the present invention provided herein.

Operation of the integrator stage 300 is essentially the same aspreviously explained with respect to FIG. 2. However, operation of theintegrator stage 300 is modified by the operation of the first andsecond switching circuits 314, 318. The first and second switchingcircuits 314, 318 receive complementary clock signals switchclk andswitchclk*. The switchclk and switchclk* signals can be generated in anyconventional manner, and typically have a lower frequency than theComp_clk signal provided to the clocked comparator 214 (FIG. 2). Theswitching circuits 314, 318 generally switch the coupling of the inputnodes to the output nodes back-and-forth in synchronicity with theswitchclk signal. As will be explained in more detail below, byperiodically switching the coupling of the input and output nodes of theswitching circuits 314, 318, and then making a determination of the datavalue stored in a memory cell by averaging multiple samples, any offsetissues with the integrator circuit 310 and l fnoise can be averaged out.

For example, assume that the differential amplifier 310 has an offsetthat causes a first offset current to flow out of differential amplifierat the node 336 (i.e., positive polarity) and a second offset current toflow into the differential amplifier at the node 338 (i.e., negativepolarity). When the switchclk signal transitions HIGH, NMOS transistors360, 361 of the first switching circuit 314 become conductive, couplingthe input node 322 to the output node 332, and coupling the input node324 to the output node 334. As for the second switching circuit 318,PMOS transistors 364, 365 become conductive, coupling the input node 336to the output node 346, and coupling the input node 338 to the outputnode 348. Thus, during the time the switchclk signal is HIGH, thepositive polarity of the first offset current adds to the output currentapplied to the capacitor 340 and the negative polarity of the secondoffset current subtracts from the output current applied to thecapacitor 342.

When the switchclk signal transitions LOW, however, the coupling of theinput and output nodes of the first and second switching circuits 314,318 switch. That is, when the switchclk signal is LOW, NMOS transistors362, 363 of the first switching circuit 314 become conductive (and NMOStransistors 360, 361 switch OFF), switching the coupling of the inputnode 322 to the output node 334 and the coupling off the input node 324to the output node 332. Similarly, in the second switching circuit 318,PMOS transistors 366, 367 become conductive (and PMOS transistors 364,365 switch OFF) switching the coupling of the input node 336 to theoutput node 348, and the coupling of the input node 338 to the outputnode 346. In this arrangement, the first offset current now adds to theoutput current applied to the capacitor 342 and the second offsetcurrent now subtracts from the output current applied to the capacitor340.

As a result of the switching of the input and output nodes of theswitching circuits 314 and 318, the positive and negative offsetcurrents are applied to each of the capacitors for an equal time. Thus,where the data state of a memory cell is based on the average ofmultiple samples taken over a period of time (preferably a multiple ofthe switchclk signal), the offset currents inherent with thedifferential amplifier 310 can be averaged out.

FIG. 4 illustrates an embodiment of a current source 400 that can besubstituted for the current sources 260, 261 illustrated in FIG. 2. Thecurrent source includes PMOS transistors 420, 422 that couple a powersupply having a voltage of Vdd to a node 426, and NMOS transistors 430,432 that couple a node 436 to ground. Each of the PMOS and NMOStransistors 420, 422, 430, and 432 have a respective voltage applied totheir gates to set the conductivity. As previously discussed, thevoltages, Vbias1, Vbias2, Vbias3, and Vbias4, are selected to set themagnitude of current supplied to the nodes 426 and 436. These voltagescan be generated and provided to the current source 400 in anyconventional manner.

The current source 400 further includes PMOS switching transistors 404a, 404 b and NMOS switching transistors 408 a, 408 b. The Down_int andUp_int signals are applied to the gates of the transistors 404 a, 408 aand 404 b, 408 b, respectively. The switching transistors 404 a, 408 a,404 b, 408 b, alternatively couple nodes 410 a, 410 b to either thepower supply or ground, depending on the logic states of the Down_intand Up_int signals. As previously discussed, the Down_int and Up_intsignals have complementary logic states, and are generated as outputsignals of the clocked comparator 214 (FIG. 2). The nodes 410 a, 410 brepresent the nodes to which the capacitors of the integrator stage 210(FIG. 2) are coupled. Thus, because of their complementary logic states,the nodes 410 a, 410 b are alternatively charged or discharged based onthe switching of the Down_int and Up_int signals.

In operation, when the Up_int signal is HIGH (and the Down_int signal isLOW), current is being supplied to the node 410 a and drawn from thenode 410 b. When the Up_int and Down_int signals switch to LOW and HIGH,respectively, current is then drawn from the node 410 a and supplied tothe node 410 b. As the Up_int and Down_int signals continue to switchlogic states, the current supplied or sunk alternates between the nodes410 a, 410 b as well.

FIG. 5 illustrates an embodiment of a clocked comparator 500 that can besubstituted for the clocked comparator 214 illustrated in FIG. 2. Theclocked comparator 500 includes a latch circuit 502 formed fromcross-coupled PMOS transistors 504, 506 and cross-coupled NMOStransistors 508, 510. A first logic state and a complementary secondlogic state are latched at nodes 514 and 516, respectively. Coupled tothe nodes 514 and 516 is an active-low set-reset (SR) flip-flop 520having two output nodes at which Up_int and Down_int signals areprovided. The Up_int and Down_int signals are provided to an averagingcircuit (not shown) for determination of the data state of a memorycell. The active-low SR flip-flop 520 is conventional in design andoperation. That is, where the logic state at the node 516 switches toLOW, the Up_int signal will be HIGH, and where the logic state at thenode 514 switches to LOW, the Down_int signal will be HIGH. Where thelogic state at both the nodes 514 and 516 are HIGH, the Up_int andDown_int signals will remain the same.

PMOS transistors 550 a, 550 b are coupled in parallel to the PMOStransistors 504 and 506, respectively. NMOS transistors 554 a, 554 b arecoupled between the cross-coupled PMOS transistors 504 and 506 and thecross-coupled NMOS transistors 508 and 510. A Comp_clk clock signal isapplied to the gates of PMOS transistors 550 a, 550 b and the NMOStransistors 554 a, 554 b. The Comp_clk signal can be produced in anyconventional manner. The clocked comparator 500 further includes NMOStransistors 560 a, 560 b coupled in parallel to the NMOS transistors 508and 510, respectively. The intout1 p and intout1 m signals are appliedto the gates of the NMOS transistors 508 and 510, respectively.

In operation, the clocked comparator 500 provides Up_int and Down_intsignals in synchronicity with the Comp_clk signal for averaging based onlogic state of the intout1 p and intout1 m signals of the integratorstage 210 (FIG. 2). Starting at the rising edge of the Comp_clk signal,the clocked comparator 500 sets the logic state of the Up_int andDown_int signals based on the logic state of the intout1 p and intout1 msignals. Upon the falling edge of the Comp_clk signal, the logic statesof the intout1 p and intout1 m signals are maintained in their presentstate until the period of the Comp_clk signal is complete.

For example, during the time the Comp_clk signal is HIGH, the latchcircuit 502 is “active,” latching logic states at the nodes 514, 516 inresponse to the logic states of the intout1 p and intout1 m signals.Note, that during the time the Comp_clk signal is HIGH, both the PMOStransistors 550 a, 550 b are OFF, thus, allowing the nodes 514, 516 tobe set according to the logic states of the intout1 p and intout1 msignals. When the latch circuit 502 is active, and the intout1 m signalis HIGH, the node 516 is pulled LOW, activating the PMOS transistor 504.This in turn pulls the node 514 HIGH and activates the NMOS transistor510. As a result, the Up_int signal provided at the output of theactive-low SR flip-flop 520 switches or remains HIGH, and the Down_intsignal switches or remains LOW. Upon the Comp_clk signal going LOW, boththe NMOS transistors 554 a, 554 b are switched OFF isolating the nodes514 and 516 from the cross coupled NMOS transistors 508 and 510.Additionally, both the PMOS transistors 550 a, 550 b become conductive,and the nodes 514, 516 are coupled to a power supply having a voltage ofVdd, pulling the nodes 514, 516 HIGH. As previously mentioned, when boththe inputs of the active-low SR flip-flop 520 are HIGH, the logic stateof the Up_int and Down_int signals are maintained in their presentstate.

When the Comp_clk signal goes HIGH again, and the logic states of theintout1 m and intout1 p signals have switched to LOW and HIGH,respectively, the node 514 will be pulled LOW and the node 516 will bepulled HIGH. As a result, the Up_int and Down_int signals will switch aswell, with the Up_int signal changing from a HIGH logic state to a LOWone, and the Down_int signal changing from a LOW logic state to a HIGHone. For the remainder of the Comp_clk cycle, the logic states of theUp_int and Down_int signals will be maintained in their present state.

It will be appreciated that the embodiments of the integrator stage 300,the current source 400, and the clocked comparator 500 shown in FIGS.3-5 and previously discussed, have been provided by way of example. Thedescription provided herein is sufficient to enable one of ordinaryskill in the art to implement the previously described circuits andprovide the same functionality and operability, but in alternativemanners. It will be further appreciated that modifications such as theseare well within the scope of the present invention.

FIG. 6 illustrates a sensing circuit 600 according to an alternativeembodiment of the present invention. The sensing circuit 600 includes afirst integrator stage 602, a second integrator stage 604, a clockedcomparator 606, and first and second switched current sources 610, 612.Operation of the sensing circuit 600 is similar to the operation of thesensing circuit 200 illustrated in FIG. 2. The sensing circuit 600 isdifferent in that a second integrator stage 604 and a second switchedcurrent source 612 has been included. The second integrator stage 604provides increased gain over the sensing circuit 200, as well as secondorder noise shaping. The integrator stage 200 can be substituted for theintegrator stages 602, 604. However, switching circuits, similar tofirst and second switching circuits 230, 234 (FIG. 2) can be omittedfrom the second integrator stage since the voltage levels of the outputsignals from the first integrator stage 602 is great enough whereinherent offsets in the second integrator stage 604 and 1/ƒ noise isless likely to cause reading errors. It will be appreciated that thedescription provided herein, including the description related to thefunction and operation of the sensing circuit 200, is sufficient toenable one of ordinary skill in the art to practice the invention.

FIG. 7 is a simplified block diagram of a memory device 700 including anMRAM array 701 having sense circuitry 710 according to an embodiment ofthe present invention. The memory device 700 further includes an addressdecoder 702 that receives addresses from external circuitry (not shown),such as a processor or memory controller, on an address bus ADDR. Inresponse to the received addresses, the address decoder 702 decodes theaddresses and applies decoded address signals to access correspondingMRAM memory cells in the MRAM array 701. A read/write circuit 704transfers data on a data bus DATA to addressed memory cells in the MRAMarray 701 during write operations, and transfers data from addressedmemory cells in the array onto the data bus during read operations. Acontrol circuit 706 applies a plurality of control signals 708 tocontrol the MRAM array 701, address decoder 702 and read/write circuit704 during operation of the MRAM 700.

In operation, the external circuitry provides address, control, and datasignals to the MRAM 700 over the respective ADDR, CONT, and DATA busses.During a write cycle, the external circuitry provides memory addresseson the ADDR bus, control signals on the CONT bus, and data on the DATAbus. In response to the control signals, the control circuit 706generates controls signals 708, to control the memory-cell array 701,address decoder 702, and read/write circuitry 704. The address decoder702 decodes the memory address on the ADDR bus and provides decodedaddress signals to select the corresponding memory cells in thememory-cell array 701. The read/write circuitry 704 receives write dataon the DATA bus, and applies the write data to the memory-cell array 701to store the data in the selected memory cells.

During a read cycle, the external circuitry provides a memory address onthe ADDR bus and control signals on the CONT bus. Once again, inresponse to the control signals, the control circuit 706 generatescontrols signals 708 to control the memory-cell array 701, addressdecoder 702, and read/write circuitry 704. In response to the memoryaddress, the address decoder 702 provides decoded address signals toaccess the corresponding memory cells in the array 701. The read/writecircuitry 704 provides data stored in the addressed memory cells ontothe DATA bus to be read by the external circuit. One skilled in the artwill understand circuitry for forming the address decoder 702,read/write circuitry 704, and control circuit 706, and thus, for thesake of brevity, these components are not described in more detail.Although only a single array 701 is shown in the MRAM 700, the MRAM mayinclude a plurality of arrays, and may also include additionalcomponents not illustrated in FIG. 7.

FIG. 8 is a block diagram of a computer system 800 including computercircuitry 802 that contains the MRAM 700 of FIG. 7. The computercircuitry 802 performs various computing functions, such as executingspecific software to perform specific calculations or tasks. Inaddition, the computer system 800 includes one or more input devices804, such as a keyboard or a mouse, coupled to the computer circuitry802 to allow an operator to interface with the computer system.Typically, the computer system 800 also includes one or more outputdevices 806 coupled to the computer circuitry 802, such output devicestypically being a printer or video display. One or more data storagedevices 808 are also typically coupled to the computer circuitry 802 tostore data or retrieve data from external storage media (not shown).Examples of typical storage devices 808 include hard and floppy disks,tape cassettes, compact disc read-only memories (CD-ROMs), read-write CDROMS (CD-RW), and digital video discs (DVDs). Moreover, although theMRAM 700 is shown as being part of the computer circuitry 802, the MRAMcan also be used as a data storage device 808 since, as previouslydescribed, the nonvolatile nature and speed of the MRAM make it anattractive alternative to other storage media devices such as harddisks.

From the foregoing it will be appreciated that, although specificembodiments of the invention have been described herein for purposes ofillustration, various modifications may be made without deviating fromthe spirit and scope of the invention. Accordingly, the invention is notlimited except as by the appended claims.

1-12. (canceled)
 13. A sensing circuit, comprising: a first integratorcircuit having a first integrator input electrically coupled to areference level, a second integrator input to which an input is applied,and first and second integrator outputs at which first and second outputsignals are provided, respectively, the integrator circuit furtherhaving an amplifier circuit having pairs of differential input andoutput nodes, the integrator circuit periodically switching theelectrical coupling of each of the differential input nodes to arespective integrator input and the electrical coupling of each of thedifferential output nodes to a respective integrator output; a secondintegrator circuit having first and second integrator inputs coupled toa respective integrator output of the first integrator circuit and firstand second integrator outputs; a comparator having first and secondinput nodes electrically coupled to a respective integrator output ofthe second integrator circuit and further having an output node, theclocked comparator periodically comparing voltage levels of the firstand second input nodes and generating an output signal having a logicstate based therefrom; and a current source having a first pair ofcurrent output nodes coupled to a respective integrator output of thefirst integrator circuit, and further having a second pair of currentoutput nodes coupled to a respective integrator output of the secondintegrator circuit, the current source switching the coupling of eachcurrent output node of the first pair to a respective integrator outputof the first integrator circuit, and each current output node of thesecond pair to a respective integrator output of the second integratorcircuit based on the logic state of the output signal of the comparator.14. The sensing circuit of claim 13 wherein the period at which thefirst integrator circuit switches the coupling of the differential inputand output nodes is greater than the period at which the comparatorcompares the voltage levels of the first and second input nodes.
 15. Thesensing circuit of claim 13 wherein the comparator comprises: a clockedlatch having a pair of complementary data nodes, first and second inputterminals coupled to a respective integrator output of the secondintegrator, the clocked latch latching data states applied to the firstand second input terminals; and a flip-flop electrically coupled to thecomplementary data nodes for providing an output signal indicative ofthe latched data states.
 16. The sensing circuit of claim 13 wherein theclocked latch latches the data states at its first and second inputterminals for a first half of the period at which the comparatorcompares voltage levels, and the clocked latch is set to a referencevoltage level for a second half of the period.
 17. The sensing circuitof claim 13 wherein the first integrator circuit comprises: a firstswitching circuit having a first pair of switches for electricallycoupling the first integrator input to the first differential input andthe second integrator input to the second differential input, and asecond pair of switches for electrically coupling the first integratorinput to the second differential input and the second integrator inputto the first differential input; and a second switching circuit having afirst pair of switches for electrically coupling the first differentialoutput to the first integrator output and the second differential outputto the second integrator output, and a second pair of switches forelectrically coupling the first differential output to the secondintegrator output and the second differential output to the firstintegrator output.
 18. The sensing circuit of claim 17 wherein the firstpairs of switches of the first and second switching circuits areactivated for the first half of the period at which the integratorcircuit switches the coupling, and the second pairs of switches of thefirst and second switching circuits are activated for the second half ofthe period.
 19. The sensing circuit of claim 13 wherein the firstintegrator circuit further comprising a feedback stage electricallycoupled to the first and second integrator outputs of the firstintegrator and the amplifier circuit, the feedback stage adjustingoutput current of the amplifier circuit based on the relative voltagelevels of the first and second integrator outputs of the firstintegrator circuit. 20-45. (canceled)